Recent international regulations governing the power quality and harmonic current pollution of the utility by users placed an increased emphasis on the problem of the interfacing electronic loads to the utility line via power circuits. By using an ac rectifier and a dc-to-dc converter for active input current shaping, it is possible to achieve unity power factor (UPF) operation out of such a dc-to-dc converter while providing regulated voltage to a load. Low frequency energy required for input-output power balance is usually stored externally in an output capacitor connected across the load. The main drawback of the input current shaper comprising the dc-to-dc converter with external energy storage is low bandwidth of the output voltage regulation limited to a few hertz. This is a consequence of the need to have a large size external output capacitor and having a single control in the dc-to-dc converter which cannot simultaneously provide both input current shaping and fast output regulation.
One known solution for achieving unity power factor operation in addition to wide bandwidth regulation of the output voltage to a load RL is shown in FIG. 1. The technique is to use a dc-to-dc converter 10, such as a forward, flyback, Cuk or Sepic converter, for output voltage regulation with an additional front-end switching regulator 11 for input current shaping, usually a boost converter coupled to an ac source 12, typically a public utility line, by a fullwave rectifier and lowpass filter 13. Low frequency energy is stored in a capacitor C1 between the two regulators. Galvanic isolation and fast output regulation are then provided by the down-stream dc-to-dc switching converter 10. This solution has several serious drawbacks: (a) power is processed twice, thus leading to lower overall conversion efficiency, (b) the complexity of such a UPF switching converter is increased and (c) both size and cost are increased.
Another class of converters combines input current shaping and fast output regulation and galvanic isolation as described in M. Madigan, R. Ericson and E. Ismail, "Integrated High Quality Rectifier-Regulators," IEEE Power Electronics Specialists Conference Proc., (1992), pp. 1043-1050; M. Brkovic and S. Cuk, "Automatic Current Shaper with Fast Output Regulation and Soft Switching," IEEE International Telecommunication Energy Conference Proc., pp. 379-378, (1993); and M. M. Jovanovic, D. M. C. Tsang and F. C. Lee, "Reduction of Voltage Stress in Integrated High-Quality Rectifier-Regulators by Variable-Frequency Control," IEEE Applied Power Electronics Conference Proc., pp. 569-575, (1994). Such automatic input current shaping with fast output regulation is shown in FIG. 2 using a Cuk converter 20 with an isolation transformer T1. A timing diagram for the operation of the converter of FIG. 2 is shown in FIG. 3. By inserting a fast diode D1 in series with an input inductor L1 of the converter 20, the inductors L1 and L2 are decoupled and a new mode of operation becomes possible. The input boost-like stage of the converter 20 comprising the inductor L1, capacitor C1 and switch Q1 is operated in a discontinuous inductor current mode (DICM), while the output buck-like stage comprising a capacitor C2, inductor L2, a diode switch D2 and a capacitor C0 is operated in a continuous conduction mode (CCM). The switch Q1 is turned ON at the beginning of a constant switching period Ts and turned OFF at the end of interval dTs, where d is duty ratio of the switch Q1 with a drive signal DRQ1 of the waveform A in FIG. 3. The diode D1 switches on automatically as the switch Q1 is switched OFF by a drive pulse DRQ1 of the waveform A in FIG. 3 at the end of the dTs. As a consequence of that new mode of operation, the voltage on the energy storage capacitor C1 is no longer uniquely determined by the input voltage to the converter 20 with the diode D1 in series and the output voltage V.sub.o to a load RL, That voltage on the capacitor C1 can be a dc voltage regardless of the input voltage waveform to the converter 20. Low-frequency energy is thus stored internally in the capacitor C1, which allows automatic current shaping and fast output voltage regulation with a single switch Q1. By keeping the interval dTs constant, which means duty ratio d constant, during a half of the line period, input current shaping is provided automatically.
The capacitor C1 of the input boost-like stage is chosen large enough to attenuate low frequency ripple component at twice the line frequency and thus provide a near constant dc voltage source for the output buck-like stage operated in CCM. Fast output regulation may then be provided by a duty ratio d for the switch Q1 that is almost constant. Variation of the duty ratio d is only required to compensate for voltage drop in parasitic resistances and provide fast output regulation. If the output buck-like stage were operated in DICM instead of in CCM, then variation in the duty ratio d would become significant for a wide range of lead changes. For instance, consider that for a step load change, the duty ratio d is also changed. Consequently, the input current waveform will change due to variation in the duty ratio d and would be distorted. That is not the case when the output buck-like stage operates in CCM because even for a no lead to full load change the duty ratio d will remain almost constant. As a consequence, the input current waveform will not be distorted.
The main disadvantage of the ac-to-dc converter in FIG. 2 and other converters described by M. Madigan, R. Ericson and E. Ismail, "Integrated High Quality Rectifier-Regulators," IEEE Power Electronics Specialists Conference Proc., (1988), pp. 334-340, is lead dependence of the voltage on the energy storage capacitor C1. This voltage is independent of the duty ratio d of the switch Q1 and therefore, cannot be controlled by the duty ratio of the switch Q1. In order to keep this voltage at an acceptable level so as to keep voltage stress on the switch Q1, diode switch D2 and capacitor C1 within an acceptable range, variable switching frequency control must be used in addition to modulation of the duty ratio d as described by M. Brkovic and S. Cuk, "Automatic Current Shaper with Fast Output Regulation and Soft Switching," IEEE International Telecommunication Energy Conference Proc., (1993), pp. 379-378; and M. M. Jovanovic, D. M. C. Tsang and F. C. Lee, "Reduction of Voltage Stress in Integrated High-Quality Rectifier-Regulators by Variable-Frequency Control," IEEE Applied Power Electronics Conference Proc., (1993), pp. 569-575. In order to accommodate variations in the load current from 5% to 100% and line voltage in a range of 80 Vrms to 270 Vrms, the switching frequency must be varied by a factor in a range from 1 to 40. Such a wide range of the switching frequency is unacceptable in practice since the implementation of the circuit at frequencies above a few hundred kHz would be extremely complex and inefficient due to circuit parasitics, mainly leakage inductance of the isolation transformer T1. Therefore, there is a strong trade-off between the voltage stress on the energy storage capacitor C1 and the switching frequency range of the control for the switch Q1. The variation in the switching frequency can be reduced if the voltage on the energy storage capacitor C1 is allowed to vary between its minimum value (which occurs at maximum load and minimum input voltage) and maximum value (which occurs at minimum load and maximum input voltage). While this approach reduces voltage stress on the capacitor C1 and switches Q1 and D1, it has negative impact on the size selected for the capacitor C1. It also requires a high current and high voltage rating for the capacitor C1.
There are three parameters which have to be taken into account in selecting the size of the energy storage capacitor C1: (a) voltage rating, (b) r.m.s. current rating and (c) minimum capacitance. The minimum voltage on the energy storage capacitor C1 is limited by the desired harmonic distortion in the input current waveform. If, for example, the sum of the voltage on the energy storage capacitor C1 and reflected output voltage NV.sub.O, which appears as the output voltage of the input boost-like stage of the converter, is kept 25% higher than the peak line voltage, the total harmonic distortion in the line current will be about 30%. At high line voltage 270 Vrms, the minimum voltage is in a range of 500 V. On the other hand, at full load and minimum line voltage 80 Vrms, the sum of the two voltages will be in a range of 150 V, as explained by M. M. Jovanovic et al., "IEEE Applied Power Electronics Conference Proc., (1993), supra.
The second parameter related to the choice of the energy storage capacitor C1 is its maximum low-frequency r.m.s. which is a combination of the low-frequency current at twice the line frequency and high-frequency current at switching frequency. The current stress in the capacitor is inversely proportional to its voltage for a given load. Consequently, the maximum current stress in the capacitor C1 is determined by the minimum line voltage with a full load condition. The current rating for an aluminum electrolytic capacitor, which is typically used for energy storage, is proportional to its capacitance.
The third parameter is minimum capacitance required to provide sufficient hold-up time. The capacitive energy storage is efficient in terms of energy density when storing energy at relatively high voltages, so significant reduction in volume of the energy storage capacitor and size of the overall power stage can be achieved.
It is thus clear that the prior-art converter of FIG. 2 and other converters described by M. Madigan et al., IEEE Power Electronics Specialists Conference Proc., (1988), supra, have in addition to variable frequency control and high voltage stress on the components, a very serious drawback which is increased size and volume of the energy storage capacitor C1 as compared to the corresponding capacitor C1 of the prior art shown in FIG. 1. The converter of FIG. 2 requires a high voltage capacitor with high current rating and large capacitance which definitely does not allow reduction in its overall size as compared to the storage capacitor C1 between the two stages 10 and 11 of the converter in FIG. 1. In addition, large variation in the storage capacitor voltage does not allow optimum design of the isolation transformer T1 or the output stage of the converter 20 in FIG. 2. Therefore, the serious drawbacks of the prior-art UPF switching converter of FIG. 2 with fast output regulation are: (a) increased voltage stress on the active and passive components, (b) need for variable switching frequency operation and (c) an oversized energy storage capacitor.